Digital frequency discriminators (DFDs) provide accurate frequencymeasurement data on pulsed and CW signals received over wide instantaneous RF bandwidths. They typically cover wide bandwidths, such as 50 to 500 MHz, 0.5 to 2 GHz, 2 to 6 GHz, 6 to 18 GHz, and 2 to 18 GHz, but are rare above 18 GHz. One of the limiting factors to achieving DFDs above 18 GHz is the problem of excess correlator phase error. What follows is a review of basic DFD capabilities and limitations, and a report on work to extend their usefulness above 18 GHz.

Figure 1 shows that DFDs come in many forms, depending on the application and performance required. DFDs are an essential building block in more complex systems, such as instantaneous-frequencymeasurement (IFM) receivers, which also include threshold circuits, an RF amplitude quantizer, RF envelope pulse-width measurement capability, and time-of-arrival (TOA) processing.

A DFD usually operates in support of a widebandwidth electronicwarfare (EW) system. Figure 2 is a simplified block diagram, showing the DFD microwave components, and reducing a complex EW system to just the essentials of an antenna, a linear RF amplifier, and a bandpass filter. DFD microwave circuits consist of a limiting RF amplifier, a phase-matched RF power divider, and (typically) a seven-correlator array, with each correlator associated with a RF delay line and delay times arranged in a binary sequence. Each microwave correlator provides both sin(θ) and cos(θ) video outputs, where θ is the relative phase between the delayed and nondelayed RF inputs to the correlator. Since the delay time associated with each microwave correlator is constant (in seconds), the relative phase between the delayed and nondelayed correlator inputs (θ) will appear to rotate as the RF input frequency is changed. The correlator output is periodic in frequency, with the period (the input frequency span required to produce 2π radians of rotation) given as:

where:
f_p = the frequency period of the correlator (in Hz) and
D = the delay time (in s).

The shortest RF delay line (identified as 1X in Fig. 2) is selected to provide just 360 deg. of phase rotation over the design unambiguous bandwidth of the DFD. The longest RF delay line (64X in Fig. 2) sets the desired RF measurement accuracy and resolution. The intermediate correlators (2X through 32X) are only present to resolve the ambiguities between the 1X and 64X correlators. The 1X through 32X correlators are provided with comparator (TTL) outputs; the 64X correlator is the only correlator employing analog video outputs. If this DFD were configured with only TTL comparators, simple decoding would produce an 8-b output data word. Using only comparators, N correlators will produce N + 1 output data bits. Using video amplifiers and digitizers on the longest delay correlator permits the expansion of frequencymeasurement resolution to 12 b or higher output resolution. The typical seven-correlator-array DFD design illustrated provides a 12-b output frequency data word; DFDs with as few as one and as many as ten correlators have been produced. Due to VSWR and other errors, the correlators are subject to phase-measurement errors. With appropriate decoding, the basic design of Fig. 2 allows each correlator to produce phase errors (relative to the adjacent correlator) to 45 deg. without causing ambiguity errors. If a similar design were to be implemented using just four correlators (1X, 4X, 16X, and 64X), the phase margin would be reduced to 22.5 deg. The problem with this abbreviated (4:1 ratio) set of correlators becomes apparent in the higher-frequency bands. The expected RMS phase error of a highfrequency correlator is approximately 6 deg. Therefore, three standard deviations is 18 deg. Over frequency and temperature this is very close to the maximum allowed phase margin of 22.5 deg. and ambiguity errors are likely.

Since RF frequency-measurement accuracy and resolution is dependent on the characteristics of the longest delay time correlator (64X, for example), it is possible to focus on the RF path including the RF preamplifier (setting the system noise figure) and the bandpass filter (setting the noise bandwidth) for performance improvements. The phase-matched power divider of Fig. 2 is replaced by a Wilkinson power divider, splitting RF inputs into delayed and nondelayed paths. The correlator simply multiplies the RF from the two paths, with the resultant video signals lowpass filtered (Fig. 3). Using this simple model, the RF input spectrum to the Wilkerson power divider is as shown in Fig. 4. The effect of the RF limiting amplifier has been temporarily ignored.

With the system model of Fig. 3 and the input spectrum of Fig. 4, it is possible to compute the measured RMS RF frequency error as a function of the RF input signal-to-noise ratio (SNR), the time delay associated with the correlator, the center of the RF passband, and the RF input bandwidth:

where:

F_e = the frequency-measurement error (in MHz RMS),
B_w = the RF bandwidth (in MHz),
B_v = the video bandwidth (in MHz),
F_o = the RF bandpass center frequency (in MHz),
D = the time delay between the RF inputs to the correlator (in µs), and
SNR = the input signal-to-noise-power ratio (dimensionless).

These relationships permits the calculation of the delay time needed to produce a desired frequency measurement accuracy over the specified RF bandwidth. Figure 5 shows the measured frequency accuracy for a typical DFD. The computation predicts a Gaussian noise distribution, even though Fig. 5 apparently shows periodic errors. These periodic errors are due to VSWR effects within the correlator's microwave circuits. In addition to the thermal-noise-based frequencymeasurement errors, three other significant error sources exist: correlator VSWR, quantization noise, and ambiguity errors.