Refinements in both the measurement probes and the calibration standard structures used with them makes it possible to perform more accurate microwave VNA S-parameter measurements on differential devices.

Larry Dangremond
Senior Product Manager For Probes and Signals
Cascade Microtech, Inc., 2430 NW 206th Ave., Beaverton, OR 97006;
(503) 601-1000, FAX: (503) 601-1002,
Internet: www.cascademicrotech.com.

Balanced or differential circuitry is becoming more commonplace in high-frequency designs as a means of reducing susceptibility to radiofrequency interference (RFI) and electromagnetic interference (EMI). Of course, testing and characterization of balanced circuits means the use of a vector network analyzer (VNA) and test set with four ports instead of two, and the study of differential and common-mode responses and mode-conversion terms. Although many test systems feature four-port test sets, single-ended measurement methods are still often used, with measurements made one port at a time and differential and common-mode characteristics calculated from the results. Especially for on-wafer testing, advances in balanced measurement techniques and their accompanying calibrations can improve the accuracy of on-wafer differential measurements.

A variety of challenges exist for onwafer or on-substrate measurements of multiport devices, especially when considering that stand-alone four-port VNA systems typically have upper-frequency limits of about 50 GHz. Some of the issues unique to f-port probebased measurements include lack of a probe-centric RF system calibration, lack of appropriate and well-designed precision probe calibration standards, and inaccuracies due to crosstalk in dual-signal probes.

Since VNA systems are commonly used for RF mixed-mode characterization, it may help to example the basic architectures of two-port (Fig. 1) and four-port (Fig. 2) VNA systems and how the different architectures impact calibration. For standard two-port measurements, the VNA’s source signal travels to the port transfer switch for routing and part of the signal is split off to the reference receiver R1. The signal connects to a device under test (DUT) at port 1 with a small amount of the reflected signal coupled and routed to the response receiver (A). Similarly, a sample of the transmitted signal is routed to a response receiver (B). At port 2, the transmitted signal continues past the coupler back to the termination path of the port transfer switch. Because this is an imperfect termination, it generates a reflection back to the DUT; a piece of this switch termination reflection signal is split off to reference receiver B. Knowledge of this switch-reflection coefficient for each choice of excitation that allows the traditional two-port 12-term calibration error model (Fig. 3) (really a 6-term model for each excitation choice) to be reduced to an 8-term error model that is used in advanced calibrations (Fig. 4).^1-3

While a four-port VNA is basically an extension of two-port architecture, there is one notable exception that impacts system calibrations. In the four-port case (Fig. 2), the R2 reference receiver in the VNA is isolated and is not able to sample any of the reflected waves. The inaccessibility of the R2 reference receiver has been an obstacle to providing a reduced error term, advanced, “probing- centric” calibration algorithm for four-port systems. Therefore, fourport VNA systems primarily have relied on a more limited resident short-open-load-through (SOLT) calibration approach. A SOLT calibration is designed to provide a coaxial reference plane using a 12-term error model. The approach relies on physical standards on substrates with previously determined, fully known electrical behavior. Probe measurements on these standards are used to extend the measurement reference plane. Unfortunately, a fundamental limitation of the 12-term error model is its sensitivity to variations in probe placement on the physical standards.⁴ Another limitation of both 12- and 8-term calibrations is that they cannot address errors due to coupling between signal paths. Four-port measurement systems that use adjacent-signal dual probes are especially susceptible.

Areas where a good deal of needed progress has been made in supporting probe-based mixed-mode measurements include reducing signal-tosignal crosstalk; improving four-port calibration structures; and developing new “probing-tolerance” calibrations for four-port systems.

Dual probes are often used for multiport mixed-mode measurements. Dual probes feature two signal lines separated by a ground or with two adjacent signal lines each with an adjacent ground. At the probe tips, the two signal lines are in close proximity and accuracy is affected by coupling between the signal lines (Fig. 5). This general coupling is typified by the simple model of capacitance between signals at the probe tip. Complicating this is that this coupling varies with DUT input impedance. Standard VNA error models cannot model this coupling. This allows for only simple isolation correction that is useful for internal VNA port leakage but does not model coupling that varies with DUT input impedance. The best option is to control crosstalk so that the uncorrected error is small.

In pursuit of this goal, dual probes and their calibration standards are designed to be as compatible as possible with available error models. To do this, both the probes and the calibration standards must exhibit high isolation. A practical goal has been to try to exceed 30 dB isolation between signals for the dual-signal probe and standard combinations.⁵ This begins to approach the behavior of state-ofthe- art single probes in the opposing configuration that often achieve 50 dB isolation between signals to 50 GHz. Conventional dual-signal probe tips have contact fingers in air or on a dielectric substrate. Since any signal path (line of sight) from signal one to signal two is blocked by a central ground, the isolation should be good. However, additional coupling possibilities are introduced when the probe is used in contact in its intended environment. The affinity for the electric field to exist in a higher dielectric material means that close to the tips of the wafer probe, the signal path is no longer line of sight but direct and through the dielectric material. Fortunately, using a microstrip architecture with inherently high isolation, it has been possible to develop dual-signal probes with high isolation.

With attention to the design of the transitions, these probes achieve better than 40 dB isolation through 40 GHz for a ground-signal-ground-signalground (GSGSG) configuration and 30 dB isolation through 40 GHz for a ground-signal-signal-ground (GSSG) configuration (Fig. 6).

For dual probes, a dual calibration standard with the appropriate physical dimensions is needed that simultaneously contacts both probe ports. For four-port systems, dual standards present a number of challenges. First, the standard itself can be a source of signal-to-signal coupling. Also, the additional conductors and the increased size of such standards (needed to contact both ports) increases the likelihood of undesired mode conversions.

Through attention to detail, it has been possible to optimize the short, open, load, and through-line (thru) calibration structures used with dual probes. For the short calibration standards, for example, when a simple bar structure is used, it can exhibit inductance in the signal-to-signal and signalto- ground paths. In this case, it is the equivalent of adding inductors to both signal paths as part of the calibration measurements. Fortunately, an improved short structure has been developed that is designed to mitigate the inductance effect (Fig. 7).

An additional problem is the unshielded contact pads on open or load standards. These unshielded pads add capacitance between the signals. An improvement involves pads that are at least partially surrounded by ground. A line-of-sight path from pad to pad that is interrupted by ground significantly reduces the pad-topad capacitance by means of electrostatic shielding.

In the design of dual probes, a coplanar waveguide (CPW) structure will have an imbalance of energy if the length of one slot is longer than another – this is compounded with dual signal probes and structures since the outside grounds are that much farther away for a given pitch. The probe grounds are also longer on the outside than in the center. This imbalance can cause mode conversions and potentially resonant behavior may be observed. An improvement has been the loop-under ground structure that connects the grounds to block the conversion of energy from coplanar to slot line mode very close to the probe tip.